The invention relates to a method for controlled application of a stator current set point value and of a torque set point value for a converter-fed rotating-field machine.
Pulse-controlled inverters with a constant input voltage are frequently used in conjunction with a field-oriented control method in order to feed rotating-field machines whose torque can be varied in a highly dynamic form to desired values within a wide rotation speed range. A drive system such as this is normally also adequate to satisfy very stringent technical demands for the control quality. Subject to the precondition of terminal currents which can be applied in any desired manner, the magnetic flux and the torque of a rotating-field machine can in principle be set continuously variably to desired values, although only if all of the electromagnetic system parameters of a sufficiently accurate description model of the machine are known. The associated terminal voltages can then be determined continuously with the aid of a single processing system—on the basis of the measurement variables of the terminal currents and the rotation speed—as a function of a nominal variable, for example for the torque. If the dynamic requirements for the control of the rotating-field machine are particularly stringent, the pulse repetition frequency must likewise be made high. This has a disadvantageous effect on the efficiency and the costs of the inverter.
Traction converters for rail vehicles do not allow high switching frequencies to be used, because of the high power density and the efficiency requirements. For example, the switching frequency in the voltage adjustment range is only in the range from 300 Hz-800 Hz for locomotives, prime movers and heavy short-distance trains and 800 Hz-2 kHz for light short-distance traffic. In addition, the available intermediate-circuit voltage must be used optimally, that is to say no voltage margin may be demanded, for control reasons. In order to avoid unacceptable network reactions, the steady-state harmonic spectrum must be defined and it must be possible to influence it. Together with the restricted switching frequency and the maximum drive capability, this requires synchronous clocking methods for the pulse-controlled inverter.
The dynamic requirements for traction converters are likewise stringent. Both the disturbance response, for example in the case of sudden overhead line voltage changes, and the drive response, for example the desired high torque dynamic range, in order to cope with sliding and skidding processes, as well as mechanical drive oscillations, must be highly dynamic in comparison to normal steady-state drives.
Furthermore, the projected maximum current load on the converter must be maintained precisely, in order to avoid overdesign of the power section. It must still be possible to apply the predetermined current by means of the control method even in the event of disturbance and reference variable changes.
A control method is therefore required in which the stator current is applied. This at the same time allows an optimum steady-state and dynamic response for presetting the torque.
Accurate and highly dynamic control of the stator current has not been possible directly until now because of the considerable harmonics caused by the clocking process and because the machine parameters (straight inductance and main inductance) are highly non-linear for harmonics.
The publication “Die stromrichternahe Antriebsregelung des Steuergerätes für Bahnautomatisierungssysteme SIBAS 32” [Controller drive control close to the converter for SIBAS 32 railroad automation systems], printed in the German Journal “eb—Elektrische Bahnen” [Electric railroads], Volume 90 (1992), Issue 11, pages 344 to 350, discloses asynchronous machine drive control close to the converter based on the field-orientation method with the major functions of measured value detection, flux model, control structure and triggering equipment.
Analogue measurement variables must be recorded for drive control based on the proven method of field orientation. Two machine currents and the input voltage of the pulse-control inverter, also referred to as the intermediate-circuit voltage, are measured. In one variant, two conductor voltages are also measured. The motor rotation speed is recorded as a further measurement variable. If an inverter is feeding two parallel-connected traction motors, then both motor rotation speeds are recorded, and the arithmetic mean value is used for control purposes.
Field-oriented control is based on knowledge of the magnitude and angle of the rotor flux. Since these variables cannot be measured directly, computer models are generally used which simulate the internal structure of the asynchronous machine. A flux model is used in order to determine the rotor flux from the measured actual values of the voltage, current and the rotation speed. This comprises two known model elements of the asynchronous machine, specifically the voltage model and the current model. At low rotation speeds, the influence of the current model dominates while, in contrast, that of the voltage model dominates at higher rotation speeds. The structure that is used thus combines the advantages of both model elements, and can be regarded as a voltage model guided by the current model. The current model includes the rotor time constant as a parameter. During operation, the rotor impedance of the machine varies to a major extent with the rotor temperature. Knowledge of the instantaneous rotor impedance is accordingly necessary for the current model to work accurately.
The central object of the signal processing system is to drive the pulse-control inverter such that the traction motor follows the required set point values. The two conductor voltages as well as the three machine currents are converted to two orthogonal components in coordinate converters. The two orthogonal current components are now converted, using the flux angle, from the stator-fixed coordinate system to a system which revolves with the rotor flux space vector, that is to say the field orientation of the current components. After filtering, the actual values of the field-forming current component and of the torque-forming current component are then produced. These current components are identical parameters at a steady-state operating point.
In order now to determine the control output variables from the nominal flux and nominal torque as reference variables, the inverse structure of the asynchronous machine is normally modeled using a so-called decoupling circuit. This calculates the required voltage components from the flux set point value, from the magnetization current set point value taken from the magnetization characteristics, from the real current set point value and from the angular velocity of the rotor flux. Two current regulators are added to the outputs of the decoupling circuit for stabilization, for the field-forming current component and the torque-forming current component.
The triggering equipment is used to match the control system to the instantaneous intermediate-circuit voltage. The drive level for the pulse width modulator is calculated from the nominal voltage and the actual value of the intermediate-circuit voltage. The task of the triggering equipment is to produce the required voltage fundamental, whose frequency and amplitude are variable, at the motor by alternate switching of the three inverter branch pairs.
The switching times are calculated using two different modulation methods, depending on the operating mode. When the frequencies and voltages are low in the starting and slow speed range, asynchronous sinusoidal modulation is used. Since a large number of switching operations in this case occur in one period of the fundamental frequency, the switching vectors and the switching angles associated with them must be determined on-line by the processor. When the ratio of the switching frequency to the fundamental frequency, the so-called pulse count, reaches a value of about 10 to 8, the inverter must be clocked in synchronism with the fundamental frequency. As the fundamental frequency rises, the limited switching frequency of the inverter means that the pulse count must be reduced in steps. Pulse patterns that have been optimized off-line are used in this case. The most important optimization criterion is the root mean square value of the harmonic current since this is the main cause of the additional losses in the motor as a result of the converter feed.
The pulse pattern must therefore be selected as a second step after optimization. In this case, a family of characteristics is created for selection of the suitable pulse system for the processor, in which the most suitable pulse pattern which satisfies the constraints of the maximum switching frequency and maintenance of the minimum pulse width and of the maximum peak current value is entered for all possible discrete values of the fundamental frequency and drive level. The pulse pattern selection level, as well as the pulse angles which have been optimized for each pulse system and for each drive level off-line, are stored in table form in the signal processor unit. The modulation type and the pulse system which is also associated with the operating point required by the control system are first of all determined from the selection level in the program model triggering equipment. The switching times in the area of the optimized patterns can be calculated as a function of the instantaneous stator frequency from the switching angles stored for the relevant drive level. In the event of pulse system changes, the times must be chosen such that no equalization processes or undesirable current spikes occur.
This so-called voltage triggering equipment allows the manipulated variable of the voltage to be predetermined just by the magnitude and angle of the fundamental, with the instantaneous values of the other electrical variables then being predetermined by the pulse pattern, such that they can no longer be influenced on-line. If the dynamics of the manipulated variable are excessive, equalization processes occur which lead to excessive torque oscillations.
The publication “Direkte Selbstregelung (DSR) für hochdynamische Drehfeldantriebe mit Stromrichterspeisung” [Direct self-regulation (DSR) for highly dynamic rotating field drives with a converter feed], printed in the German Journal “etzArchiv”, Volume 7 (1985), Issue 7, pages 211 to 218, describes direct self-regulation of a converter-fed asynchronous machine which operates without pulse width modulation, has little sensitivity to parameters and, in addition, has very good dynamic characteristics. When a rotating-field machine is being fed with a constant input voltage via a three-phase inverter, the space vector of the stator voltage can assume only seven discrete values. Ignoring the voltage which is dropped across the copper resistances of the stator windings and which is generally small in comparison to the stator voltage in the field weakening range, then the respective instantaneous value of the voltage space vector is the only factor determining the change in the velocity and the direction of the instantaneous position of the space vector for the overall flux. When the fundamental frequency clocking is in the steady state, the peak of the flux space vector therefore passes through an equilateral hexagon with a constant path velocity and with a slightly pulsating angular velocity.
In the case of fundamental frequency clocking, the only possible way to influence the torque of the asynchronous machine is to control the time intervals between the switching operations of the voltage space vector. Ignoring the voltages across the copper resistance of the stator windings (which are proportional to the current), the path velocity can very easily be reduced to the value zero in a three-phase inverter with a constant input DC voltage, specifically by connection of the seventh machine voltage space vector, whose magnitude has the value zero. As is known, any desired intermediate value of the path velocity, averaged for the pulse period, can be set by the choice of the duration of the two interval elements by using a pulse period comprising a first interval element in which the path velocity of the flux space vector is not reduced, and a second interval in which the flux space vector is stationary with respect to the stator axes.
The signal processing for direct flux self-regulation has a flux comparator and a torque comparator. An integrator is used to generate orthogonal components of the stator flux from the measured voltage values reduced by the voltage drop across the copper resistances of the stator windings in the asynchronous machine, and these components are converted to flux variables for each stator winding axis of the asynchronous machine. Each of these flux variables is compared with a flux reference variable, which can be derived from the torque control. This then results in very simple control in the basis rotation speed range in accordance with the following rule:
If the instantaneous value of the torque exceeds the set point value by more than a permissible tolerance, then the seventh space vector value whose magnitude is zero shall be applied instead of the instantaneous space vector value as determined by the flux self-regulation from the sixth outer space vector values of the machine voltage, until the actual value of the torque is below the set point value by more than the permissible tolerance. After this, the flux self-regulation once again determines the switching state of the inverter. The seventh voltage space vector value, whose magnitude is zero, can be produced, as is known, by two different switching states. Secondary conditions can be satisfied by means of appropriate selection criteria, for example the minimum switching frequency, and ensuring switching state minimum times.
This described procedure results in the angular velocity of the rotating component of the resultant flux linking, averaged over one pulse period, being automatically set to the value required to produce the desired torque, to be precise without any information about the shaft rotation speed or about instantaneous values of inductances, rotor impedance or other variables and parameters which must be known for a field-oriented control method. The slow and fast fluctuations in the input DC voltage to the inverter, which are generally always present, are taken into account automatically by the direct self-regulation, and thus have no effect on the torque, which is maintained in a predetermined tolerance band.
This direct self-regulation is suitable for a traction drive and results in an optimum dynamic response, although not in a reproducible steady-state response. Furthermore, this direct self-regulation does not permit an excessively low ratio of the switching frequency to the fundamental frequency.
The publication “Direkte Drehmomentregelung von Drehstromantrieben” [Direct torque control for three-phase drives], printed in the German Journal “ABB Technik”, No. 3, 1995, pages 19 to 24, describes newly developed direct torque control. This direct torque control [DTC] is based on the theories of field-oriented control of asynchronous machines and of direct self-regulation. In the case of direct torque control, the motor and the inverter are largely integrated. All of the switching processes of the converter are dependent on the electromagnetic state of the motor. As in the case of direct-current machines, DTC allows the flux and torque to be controlled separately. There is no need for pulse-width modulation between the motor and the inverter control.
The core units of the DTC system are the units for hysteresis control of the torque and magnetic flux as well as the logic unit for switching optimization. Another important component of the system is the accurate motor model. By means of measurements of two motor currents and of the voltage in the DC intermediate circuit, the motor model produces actual value signals for the torque, stator flux, frequency and shaft rotation speed. The set point values for the torque and flux are compared with the actual values, and the control signals are produced by two-point control of the hysteresis. The switching optimization logic determines the best voltage vector on the basis of the set point values for the torque and flux. The stator flux is controlled via the output voltage of the inverter. In the case of DTC, the stator flux and torque are kept within the hysteresis limit, that is to say within the selected tolerance band. The state set point values are changed only when the actual values of the torque and stator flux differ from their set point values by more than the permissible hysteresis. If the rotating stator flux vector reaches the upper or lower hysteresis limit, a suitable voltage vector is used to change the direction of the stator flux, and thus to keep it within the hysteresis band. The required torque is achieved by stator flux vector control.
This direct torque control also results in an optimum dynamic response, in the same way as direct self-regulation. However, the steady-state response is not reproducible, and this direct torque control does not allow an excessively low ratio of the switching frequency to the fundamental frequency, either.
The publication “Direkte Selbstregelung, ein neuartiges Regelverfahren für Traktionsantriebe im Ersteinsatz bei dieselelektrischen Lokomotiven” [Direct self-regulation, a novel control method for traction drives used for the first time in diesel-electric locomotives], printed in the German Journal “eb—Elektrische Bahnen” [Electric railroads], Volume 89, (1991), Issue 3, pages 79 to 87, describes one implementation for direct self-regulation (DSR).